I/q calibration of transmit and receive paths in ofdm fdd communication systems

ABSTRACT

I/Q gain and phase mismatches of both transmit and receive paths of an OFDM FDD transceiver are simultaneously estimated. An up-converted RF signal is generated when the transmit path performs IQ modulation on a reference signal having a single sideband tone. The up-converted RF signal is sent via a loop-back path to the receive path. A down-converted evaluation signal is generated when the receive path performs IQ demodulation on the up-converted RF signal. The single evaluation signal is used to determine the transmit path gain and phase mismatches and the receive path gain and phase mismatches. The four I/Q mismatches are estimated without using significant hardware nut otherwise used in the regular transmission of data signals. The I/Q mismatches in data signals are corrected by pre-processing the up-converted RF signals and post-processing the down-converted RF signals by adding attenuated components of the in-phase and quadrature-phase signals to each other.

BACKGROUND

1. Field

The present disclosure relates generally to wireless communicationdevices and, more specifically, to a method of estimating the I/Q gainmismatch and phase mismatch simultaneously in the transmit and receivepaths of a radio transceiver of an OFDM FDD system.

2. Background

Transceivers for wireless communication systems that use quadratureamplitude modulation typically exhibit I/Q gain and phase mismatches. Adata signal that is to be transmitted by a communication systememploying quadrature modulation is first converted into an in-phase(I-phase) transmit component and a quadrature-phase (Q-phase) transmitcomponent. Then in the analog radio transmitter, the I and Q componentsof the data signal are up-converted to a radio frequency (RF) band. Anup-converted. I-phase RF signal is generated when the I-phase transmitcomponent is mixed with an I-phase component of a local oscillatorsignal. Similarly, an up-converted. Q-phase RF signal is generated whenthe Q-phase transmit component is mixed with a Q-phase component of thelocal oscillator signal. The I-phase and Q-phase RF signals are thenadded to form a composite RF signal that is transmitted over an airinterface. Phase and gain imbalances between I and Q branches of thetransmitter are introduced when the I-phase and the Q-phase componentsof a local oscillator signal are not separated by exactly 90 degrees andwhen the amplitude of the up-converted. I-phase RF signal docs notexactly equal the amplitude of the up-converted. Q-phase RF signal. In asimilar manner, phase and gain imbalances between I and Q branches ofthe analog radio receiver are introduced when the received RF compositesignal is down-converted by mixing with the I-phase and Q-phasecomponents of a local oscillator signal. The phase and gain imbalancesbetween I and Q branches of the receiver occur when the I-phase andQ-phase components of a local oscillator signal in the receiver are notseparated by exactly 90 degrees and when the amplitude of thedown-converted I-phase signal does not exactly equal the amplitude ofthe down-converted Q-phase signal. The four impairments caused by thephase and gain imbalances in the transmitter and receiver causeinterference between positive and negative side bands of the signalspectrum.

Some wireless communications systems that are based on orthogonalfrequency-division multiplexing (OFDM) and time division duplexing (TDD)currently include functionality to estimate the I/Q gain and phaseimpairments in the transmitter and in the receiver. The existing methodsof estimating I/Q mismatches, however, have various disadvantages.First, the existing methods are performed in two steps. The transmitpath must be calibrated before the receive path can be calibrated. Thetransmit I/Q mismatches are estimated; then the transmit path iscorrected; and finally the calibrated transmit path is used to provide areference signal for estimating the I/Q mismatches of the receive path.Second, the existing methods require additional hardware that is nototherwise used in the regular transmission of data signals in currentOFDM modern transceivers, such as those used for the WLAN and WiMaxstandards. Separate dedicated hardware is needed to estimate themismatch in the transmit path. Typically this hardware is an RF envelopedetector. After the I/Q mismatches in the transmit path are estimatedusing the dedicated hardware, the local oscillator and mixers in thereceiver are used to estimate the I/Q mismatches in the receiver.

A method is sought for estimating the I/Q gain and phase impairments inboth the transmitter and the receiver of an OFDM transceiver at onetime. In addition, an apparatus is sought that can estimate the I/Q gainand phase impairments in both the transmitter and receiver of an OFDMtransceiver without requiring significant hardware not otherwise used inthe regular transmission of data signals.

SUMMARY

A method of simultaneously estimating the I/Q gain and phase mismatchesin both the transmitter and receiver of an OFDM FDD communication systemis performed using an RF loop-back path going from the output of thequadrature mixer of the transmit path to the input of the quadraturemixer of the receive path. The estimation method is performed withoutusing dedicated hardware outside of the OFDM FDD RF transceiver andbaseband modem, such as an external mixer or external envelope detector.Thus, the estimation method is performed without using hardware, otherthan a dedicated connection for the RF loop-back path, that is nototherwise used in the regular transmission of data signals. In someembodiments, even a dedicated loop-back path is not used when on-chipcoupling relays the output of the transmit path to the receive path, forexample through the substrate of the integrated circuit.

An OFDM transceiver operates in an estimation mode and in a correctionmode. In the estimation mode, the transceiver generates an evaluationsignal having a single sideband tone. An inverse Fast Fourier transformblock transforms the reference signal into in-phase (I-phase) andquadrature-phase (Q-phase) Tx components. The transmit path of thetransceiver performs IQ modulation on the I-phase and Q-phasecomponents. The I-phase Tx component is mixed with an I-phasetransmitter oscillator signal to generate an up-converted in-phasesignal. The Q-phase Tx component is mixed with a Q-phase transmitteroscillator signal to generate an up-converted quadrature-phase signal. Acomposite RF signal is generated by adding the up-converted in-phasesignal and the up-converted quadrature-phase signal. The composite RFsignal is transmitted via the RF loop-back path to the receive path ofthe transceiver, where the receive path performs IQ demodulation. Thecomposite RF signal is mixed with an I-phase receiver oscillator signalto generate a down-converted in-phase signal. The composite RF signal isalso mixed with a Q-phase receiver oscillator signal to generate adown-converted quadrature-phase signal. A Fast Fourier transform blocktransforms the down-converted in-phase signal and the down-convertedquadrature-phase signal into an evaluation signal with four sidebandtones. Each sideband tone exhibits a characteristic, such as amplitude.The transceiver uses the amplitudes of the four sideband tones tocalculate eight multiplication factors. The multiplication factors aredetermined such that when I and Q components of the transmit and receivepaths are added to each other according to the multiplication factors,the amplitude of one of the sideband tone* increases and the amplitudeof the other three sideband tones decreases. Those multiplicationfactors that entirely eliminate the gain and phase mismatches of boththe transmit and receive paths result in an evaluation signal having asingle sideband tone with the amplitudes of three of the sideband tonesbeing zero.

In the correction mode, the I/Q mismatches in data signals are correctedby pre-processing up-converted RF signals and post-processingdown-converted RF signals. In one embodiment, the transceiver iscomprised of a digital baseband integrated circuit (IC) and an analog RFtransceiver IC. The digital baseband IC includes a first correctioncircuit between the inverse Fast Fourier transform block anddigital-to-analog converters. The digital baseband IC also includes asecond correction circuit between analog-to-digital converters and theFast Fourier transform block. Each correction circuit includes fourmultipliers and two adders. Correction of the I/Q mismatches isperformed by adding attenuated components of in-phase andquadrature-phase signals to each other. For example, the secondcorrection circuit has first, second, third and fourth multipliers andfirst and second adders. The first multiplier receives a down-convertedin-phase signal and generates a first attenuated component signal. Thesecond multiplier receives a down-converted quadrature-phase signal andgenerates a second attenuated component signal. The third multiplierreceives the down-converted in-phase signal and generates a thirdattenuated component signal. The fourth multiplier receives thedown-converted quadrature-phase signal and generates a fourth attenuatedcomponent signal. The first adder generates an in-phase component signalby adding the second attenuated component signal to the first attenuatedcomponent signal. The second adder generates a quadrature-phasecomponent signal by adding the third attenuated component signal to thefourth attenuated component signal.

A Fourier transform calculation circuit generates a baseband signalusing the in-phase component signal and the quadrature-phase componentsignal. The multiplication factors used by the multipliers to generatethe attenuated signals are calculated in the estimation mode based onthe evaluation signal and are used in the correction mode to generatethe in-phase component signal and the quadrature-phase component signalthat are combined to generate the corrected baseband signal. The fourimpairments have been corrected from the corrected baseband signal. Themultiplication factors used in the correction mode are those that wouldresult in the amplitude of one sideband tone of the evaluation signalincreasing and the amplitudes of the other three sideband tones of theevaluation signal decreasing.

In another embodiment, correction circuits do not pre-process signals tobe up-converted and post-process signals that have been down-converted,but rather correct the I/Q impairments at their sources. In thisembodiment, correction multipliers and correction shifters within thequadrature mixers compensate for the gain mismatches and phasemismatches introduced by the quadrature mixers.

The foregoing is a summary and thus contains, by necessity,simplifications, generalizations and omissions of detail; consequently,those skilled in the art will appreciate that the summary isillustrative only and does not purport to be limiting in any way. Otheraspects, inventive features, and advantages of the devices and/orprocesses described herein, as defined solely by the claims, will becomeapparent in the non-limiting detailed description set forth herein.

BRIEF DESCRIPTION OF THE DRAWINGS

Like numerals indicate like components in the accompanying drawings ofthe various embodiments.

FIG. 1 is a simplified schematic block diagram of an OFDM transceiverthat performs a method of simultaneously estimating the gain and phasemismatches of both the transmitter and the receiver;

FIG. 2 is a flowchart of steps for estimating and correcting the gainand phase mismatches of the transceiver of FIG. 1;

FIG. 3 is a diagram in the frequency domain showing a reference signalgenerated by the transceiver of FIG. 1;

FIG. 4 is a complex equation describing the I-phase and Q-phasecomponents output by an inverse Fast Fourier transform (IFFT) block ofthe transceiver of FIG. 1;

FIG. 5 shows equations describing the output of in-phase andquadrature-phase transmit mixers of the transceiver of FIG. 1;

FIG. 6 is an equation describing an up-convened in-phase signalgenerated by mixing the I and Q components output by the IFFT block withthe real part of the output of the in-phase transmit mixer of thetransceiver of FIG. 1;

FIG. 7 shows equations describing the output of the receive mixers ofthe transceiver of FIG. 1 based solely on the local oscillator signalsbefore mixing with the composite RF signal;

FIG. 8 shows equations describing the convolution of the output of thereceive mixers with the characteristics of the RF loop-back path of thetransceiver of FIG. 1;

FIG. 9 is an equation representing the baseband signal that results whenthe composite RF signal is mixed with the local oscillator signal of thereceiver of the transceiver of FIG. 1;

FIG. 10 is a diagram in the frequency domain showing the magnitudes ofthe four complex frequency coefficients of the evaluation signalgenerated using the reference signal;

FIG. 11 is an equation representing the complex frequency coefficientsof FIG. 10 as a function of the gain and phase mismatches of thetransmitter and receiver and of the time delay of the RF loop-back path;

FIG. 12 shows equations describing the relationship between gaincomponents and the gain mismatches;

FIG. 13 shows equations describing the relationship between the phasecomponents and the phase mismatches;

FIG. 14 shows equations describing the real and imaginary parts of thecomplex amplitudes described in the equation of FIG. 11;

FIG. 15 shows equations representing the gain and phase components as alinear combination of the real and imaginary parts of the complexamplitudes described in the equation of FIG. 11;

FIG. 16 shows equations expressing the gain components in terms of thecomplex frequency coefficients that were measured from the evaluationsignal 105;

FIG. 17 shows equations describing the gain impairment of thetransmitter and the gain impairment of the receiver in terms of the gaincomponents;

FIG. 18 shows equations expressing the phase components in terms of thecomplex frequency coefficients that were measured from the evaluationsignal 105;

FIG. 19 is an equation representing the baseband components of a complexdata signal after up-conversion and down-conversion.

FIG. 20 shows the coefficients of the inverse matrix that correspond tothe multiplication factors of the multipliers in correction circuits ofthe transceiver of FIG. 1;

FIG. 21 is an equation expressing the coefficients of the Q matrix as afunction of the gain and phase mismatches of the transmitter of thetransceiver of FIG. 1;

FIG. 22 is an equation expressing the multiplication factors of theinverse matrix as a function of the gain and phase mismatches of thetransmitter of the transceiver of FIG. 1;

FIG. 23 is an equation expressing the coefficients of the Q matrix as afunction of the gain and phase mismatches of the receiver of thetransceiver of FIG. 1;

FIG. 24 is an equation expressing the multiplication factors of theinverse matrix as a function of the gain and phase mismatches of thereceiver of the transceiver of FIG. 1;

FIG. 25 is a schematic block diagram of another embodiment oftransceiver of FIG. 1 in which the correction circuits are located in ananalog integrated circuit; and

FIG. 26 is a schematic block diagram of yet another embodiment of thetransceiver of FIG. 1 in which the gain and phase impairments are notpre-processed and post-processed by correction circuits, but rather arecorrected at their sources in the quadrature mixers.

DETAILED DESCRIPTION

Reference will now be made in detail to the various embodiments,examples of which are illustrated in the accompanying drawings.

FIG. 1 is a simplified block diagram of a transceiver 10 that performsIQ modulation and demodulation for a wireless communication system thatis based on orthogonal frequency division multiplexing (OFDM) usingfrequency division duplexing (FDD). Transceiver 10 includes an RFloop-back path 11 from the quadrature mixer of the transmit path to thequadrature mixer of the receive path. RF loop-back path 11 is used toperform a method of simultaneously estimating the gain and phasemismatches in the in-phase and quadrature-phase paths of both thetransmitter and the receiver. The I/Q calibration method is performedwithout using any additional hardware external to transceiver 10. Themethod of simultaneously estimating the I/Q gain and phase mismatches(impairments) in both the transmitter and receiver is particularlysuited for FDD modern transceivers because these receivers independentlygenerate the transmit and receive frequencies. Moreover, the method isparticularly suited for OFDM modern transceivers because OFDM receiversinclude signal generation and processing devices capable of performingthe I/Q mismatch estimation of the method. Thus, devices that operateunder the following wireless communication standards can perform themethod without significant additional hardware: 3GPP Long-Term Evolution(LTE), Ultra Mobile Broadband (UMB) also known as Evolution-DataOptimized Revision C, and FDD WiMax (IEEE 802.16). In one example,transceiver 10 is part of a mobile communication device, such as a cellphone or a personal digital assistant (PDA). In another example,transceiver 10 is part of a base station that receives signals from andtransmits signals to mobile communication devices.

Transceiver 10 includes RF loop-back path 11, a transmitter 12, areceiver 13, a digital signal processor (DSP) 14, a serial peripheralinterface (SPI) bus 15 and an I/Q calibration register 16. Transceiver10 is coupled to a duplexer 17 and an antenna 18. Transmitter 12performs IQ modulation based on OFDM, and receiver 13 performs IQdemodulation based on OFDM. In one embodiment, the functions oftransmitter 12 and receiver 13 are performed on both an analog radiofrequency (RF) transceiver integrated circuit (IC) 19 and on a digitalbaseband IC 20. RF transceiver IC 19 and digital baseband IC 20communicate over SPI serial bus 15. I/Q calibration register 16 can bewritten to from SPI serial bus 15 and is used to control RF loop-backpath 11. In some embodiments, SPI serial bus 15 is replaced with aparallel interface between RF transceiver IC 19 and digital baseband IC20. Communication between IC 19 and IC 20 is faster over the parallelinterface than over a single pin interface, such as SPI serial bus 15.

In another embodiment of transceiver 10 not shown in FIG. 1, both theanalog and the digital functions of transceiver 10 are performed on asingle integrated circuit, called a system on a chip (SOC). The systemon a chip includes the quadrature mixers of the transmit and receivepaths, as well as baseband processing and digital control blocks. Thebaseband processing block performs the calculations of the I/Qcalibration method, and the digital control block controls the transmitand receive operations to correct for the I/Q gain and phase mismatchesin the transmitter and receiver.

In the embodiment of FIG. 1, transmitter 12 includes an inverse FastFourier transform (IFFT) block 21, a first correction circuit 22, afirst digital-to-analog converter (DAC) 23, a second DAC 24, a first lowpass filter 25, a second low pass filter 26, a transmit local oscillator27, a transmit frequency synthesizer 28, a transmit phase shifter 29, anin-phase transmit mixer 30, a quadrature-phase transmit mixer 31, asummer 32 and a power amplifier 33. Receiver 13 includes a low-noiseamplifier 34, a receive local oscillator 35, a receive frequencysynthesizer 36, a receive phase shifter 37, an in-phase receive mixer38, a quadrature-phase receive mixer 39, a third low pass filter 40, afourth low pass filter 41, a first analog-to-digital converter (ADC) 42,a second ADC 43, a second correction circuit 44 and a Fast Fouriertransform (FFT) block 45. Digital baseband IC 20 tunes transmitter 12and receiver 13 by controlling the frequency ω_(a) of a local oscillator(LO) signal 46 supplied by transmit frequency synthesizer 28 to mixers30-31 and the frequency ω_(b) of a LO signal 47 supplied by receivefrequency synthesizer 36 to mixers 38-39. Transmit local oscillator 27generates an oscillating signal, and transmit frequency synthesizer 28adjusts the oscillating signal to generate LO signal 46. Similarly,receive frequency synthesizer 36 conditions the oscillating signaloutput by receive local oscillator 35 to generate LO signal 47.

In the method of simultaneously estimating both receive and transmit I/Qmismatches, transceiver 10 uses the frequency offset between thetransmitter and the receiver of the FDD communication system to decouplethe determination of the transmitter I/Q mismatches from thedetermination of the receiver I/O mismatches and thereby rendersdetermination of the four mismatches resolvable in one set ofcalculations. Transceiver 10 estimates both the I/Q gain and phasemismatches in the transmitter and the I/Q gain and phase mismatches inthe receiver at the same time instead of first estimating the I/Qmismatches of the transmitter, then calibrating the transmit path, andthen estimating the I/Q mismatches of the receiver.

Where the transmitter and the receiver use the same frequency, such asin time division duplexing (TDD) systems, the I/Q mismatches of thetransmitter cannot be resolved separately from the I/Q mismatches of thereceiver without using additional hardware. Thus. I/Q estimation methodsfor TDD systems use additional hardware to downconvert the RF transmitsignal at a different frequency than that at which the receiverdownconverts the RF transmit signal. Conventional I/Q estimation methodsfor both TDD and FDD systems have first calibrated the transmit pathbefore estimating the I Q mismatches of the receive path. Because themethod of simultaneously estimating both receive and transmit I Qmismatches performs the processing for all four I Q mismatch estimationsat one time, the estimation method is at least twice as fast asconventional I Q estimation methods.

FIG. 2 is a flowchart showing steps of a method 48 by which all four I/Qimpairments of the transmit and receive paths of an OFDM transceiver areestimated at the same time and corrected. The operation of transceiver10, as shown in FIG. 1, is explained in detail in connection with steps49-61 listed in FIG. 2. Transceiver 10 operates in two modes: anestimation mode and a correction mode. The estimation mode correspondsto a first phase of method 48 in which the four I/Q impairments areestimated. The correction mode corresponds to a second phase of method48 in which the four I/Q impairments are corrected using firstcorrection circuit 22 and second correction circuit 44. In a first step49, the method 48 of simultaneously estimating both receive and transmitI/Q mismatches uses resources in DSP 14 to generate a reference signal62. Reference signal 62 is generated in the frequency domain and has asingle sideband tone.

FIG. 3 is a diagram in the frequency domain showing the amplitudes ofvarious “frequency bins” of reference signal 62. In one exemplaryimplementation, there are sixty-four frequency bins. FIG. 3 illustratesthat reference signal 62 has an amplitude in only the frequency bin 63at frequency ω₀. The amplitudes of the frequency bins of referencesignal 62 are represented in FIG. 1 as (0, 0, 1, 0 . . . 0, 0, 0). Thus,reference signal 62 has a single sideband tone at ω₀.

In step 50, inverse Fast Fourier transform block 21 transforms referencesignal 62 from the frequency domain into the time domain. IFFT 21outputs the time domain transform of reference signal 62 in the form ofa real I-phase Tx component 64 and an imaginary Q-phase Tx component 65.I-phase Tx component 64 can be represented as cos(ω₀t), and Q-phase Txcomponent 65 can be represented as jsin(ω₀t).

FIG. 4 is a complex equation 66 describing I-phase component 64 andQ-phase component 65 output by IFFT 21. Equation 66 described components64 and 65 in the time domain as x(t) in terms of ω₀, where j is theimaginary unit.

I-phase Tx component 64 and Q-phase Tx component 65 are received byfirst correction circuit 22. First correction circuit 22 includes fourmultipliers 67-70 and two adders 71-72. I-phase Tx component 64 isreceived by first multiplier 67 and third multiplier 69, and Q-phase Txcomponent 65 is received by second multiplier 68 and fourth multiplier70. First adder 71 receives the output of first multiplier 67 and secondmultiplier 68, while second adder 72 receives the output of thirdmultiplier 69 and fourth multiplier 70. In the first phase of method 48in which the four I/Q impairments are estimated and before theimpairments are corrected, first correction circuit 22 merely passes onI-phase component 64 and Q-phase component 65 unchanged todigital-to-analog converters 23-24. In the first phase while transceiver10 is operating in the estimation mode, the multiplication factors ofsecond and third multipliers 68-69 are set to zero. First DAC 23receives I-phase Tx component 64 and outputs an analog I-phase Txcomponent. Second DAC 24 receives Q-phase Tx component 65 and outputs ananalog Q-phase Tx component. The analog Tx components are output bydigital baseband IC 20 and received by analog RF transceiver IC 19. Theanalog I-phase Tx component is Filtered by first low pass filter 25, andthe analog Q-phase Tx component is filtered by second low pass filter26.

In a step 51, the converted and filtered I-phase Tx component isup-converted by mixing it with transmitter LO signal 46. Transmitfrequency synthesizer 28 generates transmitter LO signal 46 with afrequency ω_(a) and provides in-phase transmitter LO signal 73 toin-phase transmit mixer 30. In-phase transmit mixer 30 generates anup-converted in-phase signal 74. In a step 52, the converted andfiltered Q-phase Tx component is up-converted by quadrature-phasetransmit mixer 31. Transmit phase shifter 29 receives transmitter LOsignal 46, delays the phase by 90 degrees and outputs a quadrature-phasetransmitter LO signal 75. Mixer 31 mixes quadrature-phase transmitter LOsignal 75 with the converted and filtered Q-phase Tx component andoutputs an up-converted quadrature-phase signal 76. In FIG. 1, transmitphase shifter 29 is depicted as shifting the phase of transmitter LOsignal 46 by “90°−φ_(a)/2”. In addition, in-phase transmitter LO signal73 is depicted as having a phase that is shifted “+φ_(a)/2” fromtransmitter LO signal 46. The φ_(a) represents the phase impairment bywhich up-converted quadrature-phase signal 76 is not shifted exactly 90degrees compared to up-converted in-phase signal 74. Similarly, amultiplier 77 that multiplies by “1+ε_(a) 2” and a multiplier 78 thatmultiplies by “1−ε_(a) 2” represent the gain mismatch between thein-phase and quadrature-phase paths of the transmitter. The ε_(a)represents the gain impairment by which the amplitude of up-convertedin-phase signal 74 is greater than the amplitude of up-convertedquadrature-phase signal 76.

FIG. 5 shows an equation 79 describing the output of in-phase transmitmixer 30 based solely on in-phase transmitter LO signal 73 and beforemixing with the converted and filtered I-phase Tx component obtainedfrom the real I-phase Tx component 64. FIG. 5 also shows an equation 80describing the output of quadrature-phase transmit mixer 31 based solelyon quadrature-phase transmitter LO signal 75 and before mixing with theconverted and filtered Q-phase Tx component obtained from the imaginaryQ-phase Tx component 65. Equation 79 describes the real part andequation 80 describes the imaginary part of the output of the transmitmixers as a function of the frequency ω_(a) of transmitter LO signal 46,the phase mismatch φ_(a) and the gain mismatch ε_(a).

In a step 53, summer 32 adds up-converted in-phase signal 74 andup-converted quadrature-phase signal 76 and outputs a composite RFsignal 81. In the estimation mode, no signals are conveyed throughantenna duplexer 17 and transmitted by antenna 18. In the estimationmode, composite RF signal 81 is conveyed via RF loop-back path 11 fromthe quadrature mixer of transmitter 12 to the quadrature mixer ofreceiver 13. In the first phase of method 48, a switch 82 in RFloop-back path 11 is closed. In the second phase of method 48, switch 82is open. In one embodiment, switch 82 is comprised of two transistors.Switch 82 is closed when a switching signal 83 is asserted. In oneembodiment, switching signal 83 is asserted when a digital one iswritten into the sixth bit of I/Q calibration register 16. Digitalbaseband IC 20 controls switch 82 by communicating across SPI serial bus15 and writing the digital one into the sixth bit of I/Q calibrationregister 16.

In another embodiment, no dedicated loop-back path is used to conveycomposite RF signal 81 from the quadrature mixer of transmitter 12 tothe quadrature mixer of receiver 13. Instead, the substrate of IC 19 andIC 20 or the substrate of the system on a chip (SOC) acts as the RFloop-back path, and composite RF signal 81 is conveyed between thetransmitter and receiver through substrate coupling. Alternatively,coupling of signals from transmitter 12 to receiver 13 can be performedthrough transformers. In yet another embodiment, an off-chip coupler isused to convey composite RF signal 81 from the quadrature mixer oftransmitter 12 to the quadrature mixer of receiver 13.

FIG. 6 shows an equation 84 describing the up-converted output of thequadrature mixer of transmitter 12. Equation 84 describes bothup-converted in-phase signal 74 and up-converted quadrature-phase signal76. Up-converted in-phase signal 74 is the product of equation 66describing the I and Q components 64-65 output by IFFT 21 multiplied byequation 79 describing the real part of the output of in-phase transmitmixer 30 generated with in-phase transmitter LO signal 73. Up-convertedquadrature-phase signal 76 is the product of equation 66 describing theI and Q components 64-65 multiplied by equation 80 describing theimaginary part of the output of quadrature-phase transmit mixer 31generated with quadrature-phase transmitter LO signal 75. Equation 84describes up-converted signal 74 and 76 as the function

{x(t)·a(t)} of the frequency ω₀ of the single sideband tone of referencesignal 62, the frequency ω_(a) of transmitter LO signal 46, the transmitphase mismatch φ_(a) and the transmit gain mismatch ε_(a).

In a step 54, composite RF signal 81 is down-converted by mixing it withreceiver LO signal 47. Receive frequency synthesizer 36 generatesreceiver LO signal 47 having a frequency ω_(b) and provides an in-phasereceiver LO signal 85 to in-phase receive mixer 38. In-phase receivemixer 38 generates a down-converted in-phase signal 86. In a step 55,composite RF signal 81 is also down-converted by mixing it with aquadrature-phase receive LO signal 87. Receive phase shifter 37 receivesreceiver LO signal 47, delays the phase by 90 degrees and outputsquadrature-phase receive LO signal 87. Mixer 39 mixes quadrature-phasereceive LO signal 87 with composite RF signal 81 and outputs adown-converted quadrature-phase signal 88. In FIG. 1, receive phaseshifter 37 is depicted as shifting the phase of receiver LO signal 47 by“90°−φ_(b)/2”. In addition, in-phase receive LO signal 85 is depicted ashaving a phase that is shifted “+φ_(b)/2” from receiver LO signal 47.The φ_(b) represents the phase impairment by which down-convenedquadrature-phase signal 88 is not shifted exactly 90 degrees compared todown-converted in-phase signal 86. Similarly, a multiplier 89 thatmultiplies by “1+ε_(b)/2” and a multiplier 90 that multiplies by“1−ε_(b)2” represent the gain mismatch between the in-phase andquadrature-phase paths of the receiver. The ε_(b) represents the gainimpairment by which the amplitude of down-converted in-phase signal 86is greater than the amplitude of down-converted quadrature-phase signal88.

FIG. 7 shows an equation 91 describing the output of in-phase receivemixer 38 based solely on in-phase receiver LO signal 85 and beforemixing with composite RF signal 81. FIG. 7 also shows an equation 92describing the output of quadrature-phase receive mixer 39 based solelyon quadrature-phase receiver LO signal 87 and before mixing withcomposite RF signal 81. Equation 91 describes the real part and equation92 describes the imaginary part of the output of the quadrature mixer ofreceiver 13 as a function of the frequency ω_(b) of receiver LO signal47, the phase mismatch φ_(b) and the gain mismatch ε_(b).

When characterizing composite RF signal 81 that is down-converted in thequadrature mixer of receiver 13 for purposes of estimating I/Qmismatches, a more accurate estimation is obtained by considering theattenuation and delay in composite RF signal 81 introduced by RFloop-back path 11. RF loop-back path 11 introduces unknown gain, phaseand delay errors. The RF loop-back path 11 causes gain scaling and atime delay across the channel connection from transmitter 12 to receiver13. The characteristics of the channel connection are described by theequation:

c(t)=β·δ(t−τ)  (93)

where β represents the gain scaling, δ represents the phase shift, and τrepresents the time delay. The function δ(t) denotes the Dirac impulse.In order more accurately to describe down-converted in-phase signal 86and down-converted quadrature-phase signal 88, the output of the receivemixers 38-39 is first convolved with the characteristics of the channelconnection before the product of composite RF signal 81 and receiver LOsignal 47 is calculated.

FIG. 8 shows equations 94-95 that describe the convolution of the outputof the receive mixers 38-39 with the characteristics of RF loop-backpath 11. Equation 94 shows

¦c(t)*b(t)¦ and represents the convolution of equation 93 with the realpart of the output of the quadrature mixer of receiver 13 described byequation 91. Equation 95 shows

¦c(t)*b(t)¦ and represents the convolution of equation 93 with theimaginary part described in equation 92.

FIG. 9 shows an equation 96 representing the baseband signal thatresults when composite RF signal 81 is mixed with receiver LO signal 47and down-converted. Equation 96 is the product of

¦x(t)·a(t)¦ (equation 84 representing the real part of the up-convertedin-phase signal 74) and ¦c(t)·b(t)¦ (equations 94-95 representing thereal and imaginary parts of the convolution of the receive mixers 38-39with RF loop-back path 11). Thus, equation 96 is the product

¦x(t)·a(t)¦·¦b(t)·c(t)¦. In order to simplify the calculation of the I Qmismatches, equation 96 considers only the product of up-convertedin-phase signal 74 and the convolution of the characteristics of mixers38-39 with RF loop-back path 11 that falls within a baseband bandwidth.Signal components of the multiplication of equation 84 and equations94-95 that fall at the frequency (ω_(a)+ω_(b)) are ignored because thefrequency (ω_(a)+ω_(b)) is roughly twice the carrier frequency ofcomposite RF signal 21 and are assumed to be filtered out by third lowpass filter 40 and fourth low pass filter 41.

Down-converted in-phase signal 86 is filtered by third low pass filter40, and down-converted quadrature-phase signal 88 is filtered by fourthlow pass filter 41. RF transceiver IC 19 then passes the filtereddown-converted signals 86 and 88 to digital baseband IC 20. Firstanalog-to-digital converter (ADC) 42 digitizes the filtereddown-converted in-phase signal 86, and second ADC 43 digitizes thefiltered down-converted quadrature-phase signal 88.

The digitized and filtered down-convened signals 86 and 88 are receivedby second correction circuit 44. Second correction circuit 44 includesfour multipliers 97-100 and two adders 101-102. Digitized and filteredin-phase signal 86 is received by fifth multiplier 97 and seventhmultiplier 99, and digitized and filtered quadrature-phase signal 88 isreceived by sixth multiplier 98 and eighth multiplier 100. Third adder101 receives the output of fifth multiplier 97 and sixth multiplier 98,while fourth adder 102 receives the output of seventh multiplier 99 andeighth multiplier 100. In the first phase of method 48 in which the fourI/Q mismatches are estimated and before the mismatches are corrected,second correction circuit 44 merely passes on digitized and filteredin-phase signal 86 and digitized and filtered quadrature-phase signal 88unchanged to Fast Fourier transform (FFT) block 45. In the first phasewhile transceiver 10 is operating in the estimation mode, themultiplication factors of sixth and seventh multipliers 98-99 are set tozero. In the estimation mode, second correction circuit 44 outputs anI-phase Rx component 103 in substantially the same form as the digitizedand filtered in-phase signal 86 that second correction circuit 44receives from ADC 42. In the estimation mode, second correction circuit44 outputs a Q-phase Rx component 104 in substantially the same form asthe digitized and filtered quadrature-phase signal 88 that secondcorrection circuit 44 receives from ADC 43.

In a step 56, FFT block 45 receives the digitized and filtereddown-converted signals 86 and 88 and transforms them into an evaluationsignal 105. Down-converted in-phase signal 86 and down-convertedquadrature-phase signal 88 are received in the time domain by FFT block45, and FFT block 45 outputs evaluation signal 105 in the frequencydomain. Whereas reference signal 62 has a single sideband tone,evaluation signal 105 has four sideband tones. Each of the four sidebandtones exhibits an amplitude, a phase and other characteristics. Thecharacteristics of the sideband tones are represented by the complexfrequency coefficients C₊₁, C₊₂, C⁻¹ and C⁻² that correspond to thevarious frequency bins (0, 0 . . . C₊₁ . . . C₊₂ . . . C⁻¹ . . . C⁻² . .. 0, 0, 0) indicated in FIG. 1. Assuming that ω_(a)>ω_(b) andω₀<(ω_(a)−ω_(b)), then the four sideband tones with the characteristicsin the magnitudes defined by the complex Fourier coefficients C₊₁, C₊₂,C⁻¹ and C⁻² fall at the frequencies (ω_(a)−ω_(b)+ω₀), (ω_(a)−ω_(b)−ω₀)−(ω_(a)−ω_(b)+ω₀) and −(ω_(a)−ω_(b)−ω₀), respectively.

FIG. 10 is a diagram of evaluation signal 105 in the frequency domainshowing the magnitudes of the complex frequency coefficients C⁻¹, C⁻²,C⁻¹ and C⁻² at the four corresponding frequency bins. In the estimationmode, the magnitudes of the complex frequency coefficients aredetermined by measuring evaluation signal 105.

In a step 57, DSP 14 determines the first characteristic C⁻¹ of thefirst sideband tone, the second characteristic C⁻² of the secondsideband tone, the third characteristic C⁻¹ of the third sideband toneand the fourth characteristic C⁻² of the fourth sideband tone. Themethod 48 of estimating all four I/Q impairments at the same time usesthe complex amplitudes C⁻¹, C⁻², C⁻¹ and C⁻² of the sidebands ofevaluation signal 105 to calculate the multiplication factors for theeight multipliers of correction circuits 22 and 44. The multiplicationfactors correspond to the amplitudes of the I-phase and Q-phase inputsto correction circuits 22 and 44 that can be added to each other suchthat one of the complex amplitudes C⁻¹, C⁻², C⁻¹ and C⁻² increases andthe other three complex amplitudes decrease. The increase of one complexamplitude and the decrease of the other three complex amplitudescorresponds to a reduction in the four I/Q impairments.

FIG. 11 shows an equation 106 representing the magnitude of complexfrequency coefficients C⁻¹, C⁻², C⁻¹ and C⁻² as a function of the gainand phase mismatches ε_(a) and φ_(a) of transmitter 12, the gain andphase mismatches ε_(b) and φ_(b) of receiver 13 and the time delay τ ofRF loop-back path 11. The determination of the multiplication factorsfor the eight multipliers of correction circuits 22 and 44 is performedin several calculations using the observed complex amplitudes C⁻¹, C⁻²,C⁻¹ and C⁻² of the sidebands of evaluation signal 105. First, equation106 is rewritten in terms of gain components K₁₁, K₁, and K₁ and K andphase components φ₊ and φ⁻. Moreover, the time delay t is expressed interms of γ, where γ=ω_(b)·τ. FIG. 12 shows equations describing therelationship between the gain components K₊₊, K⁺⁻, K⁻⁺ and K⁻ and thegain mismatches ε_(a) and ε_(b). FIG. 13 shows equations describing therelationship between the phase components φ₊ and φ⁻ and phase mismatchesφ_(a) and φ_(b).

Then, the complex amplitudes described in equation 106 are divided intotheir real and imaginary parts. FIG. 14 shows an equation 107 describingthe real pan of the complex amplitudes of equation 106. FIG. 14 alsoshows an equation 108 describing the imaginary part of the complexamplitudes of equation 106. Equation 107 is then rewritten to expressthe gain components K₁, K₁₁₁, K₁ and K₁ and phase components φ₁ and φ₁as a linear combination of the real part of the complex amplitudes C⁻¹,C⁻², C⁻¹ and C⁻². FIG. 15 shows an equation 109 representing the gainand phase components as a linear combination of the real pan of thecomplex amplitudes. FIG. 15 also shows an equation 110 representing thegain and phase components as a linear combination of the imaginary partof the complex amplitudes.

The equations 109 and 110 are then used to solve for the gain componentsK₁, K₁, K₁ and K₁. The gain component of each row of equations 109 and110 is calculated as K=(

row¦²+

¦row¦²)′². FIG. 16 shows equations 111-114 that express the gaincomponents K₁, K₁, K₁ and K₁ in terms of the complex frequencycoefficients C₁₁, C₁₂, C₁₁ and C₁₂ that were measured from evaluationsignal 105. The values of the gain components are calculated using themeasured values of the complex frequency coefficients.

Next, the gain impairment ε_(a) of the quadrature mixer of transmitter12 and the gain impairment ε_(b) of the quadrature mixer of receiver 13are determined by solving the equations of FIG. 12 for ε_(a) and ε_(b)and inserting the values obtained from equations 111-114 for the gaincomponents K₁, K₁, K₁ and K₁. The gain components K₁, K₁, K₁, and K₁were expressed in equations 111-114 in terms of the complex amplitudesC₁₁, C₁₂, C₁₁ and C₁₂. Thus, the gain impairments ε_(a) and ε_(b) aredetermined from the observed complex amplitudes C₊₁, C₊₂, C⁻¹ and C⁻².

FIG. 17 shows two equations 115-116 that express the gain impairmentε_(a) in terms of the gain components K₊₊, K⁺⁻, K⁻⁺ and K⁻¹. The resultsof the two equations 115-116 will likely not be identical because ofnoise introduced by transceiver 10. Thus, the two values of the gainimpairment ε_(a) in transmitter 12 are determined using both equations,and the results are averaged. FIG. 17 also shows two equations 117-118that express the gain impairment ε_(b) in receiver 13 in terms of thegain components K₊₊, K⁺⁻, K⁻⁺ and K⁻⁻. The results of the two equations117-118 are also averaged to obtain the gain impairment ε_(b). Note thatthe values for ε_(a) and ε_(b) in equations 115-118 are no longerdependent on the gain of the channel connection represented by the gainscaling factor β, which is present in equations 111-114.

Next, the phase mismatch φ_(a) of the quadrature mixer of transmitter 12and the phase mismatch φ_(b) of the quadrature mixer of receiver 13 aredetermined. Using the equations of FIG. 13, the phase mismatches φ_(a)and φ_(b) are expressed in terms of the phase components φ₁ and φ₁ asfollows:

φ_(a)=φ₁+φ₁  (119)

φ_(a)=φ₁−φ₁  (120).

Then, the phase components φ₁ and φ₁ are extracted from equations 109and 110 of FIG. 15. Each of the four rows of equation 110 is divided bythe corresponding row of equation 109. The phase is then extracted byapplying tan⁻¹. Then, the rotation introduced by the channel connectionis removed by adding rows one and four and rows two and three. FIG. 18shows equations 121 and 122 that express the phase components φ₁ and φ₁in terms of the complex frequency coefficients C₁₁, C₁₂, C₁₁ and C₁₂.Thus, the phase mismatches φ_(a) and φ_(b) are determined from theobserved complex amplitudes C₁₁, C₁₂, C₁₁ and C₁₂.

In the last step of the first phase of method 48 and while transceiver10 is still operating in the estimation mode, the multiplication factorsto be used in the second phase are determined. Then in the second phaseof method 48 while transceiver 10 is operating in the correction mode,the multipliers 67-70 and 97-100 are set using the multiplicationfactors in order to correct for the four I1Q impairments (gainimpairments ε_(a) and ε_(b) and phase mismatches φ_(a) and φ_(b)).

In a step 58, the amplitudes of I-phase Tx component 64 and Q-phase Txcomponent 65 that are added to each other to correct for the gain andphase impairments ε_(a) and φ_(a) are determined. For example, anamplitude of I-phase Tx component 64 as governed by the multiplicationfactor of multiplier 67 and an amplitude of Q-phase Tx component 65 asgoverned by the multiplication factor of multiplier 68 are added suchthat two of the frequency coefficients (C₊₁ and C⁻¹) increase and theother two coefficients (C₊₂ and C⁻²) decrease. (The correction of thegain and phase impairments ε_(b) and φ_(b) of the receive path then alsocauses the frequency coefficient C₊₁ to decrease.) The increase in afrequency coefficient represents an increase in a characteristic of thecorresponding sideband tone of evaluation signal 105, such as theamplitude of the sideband tone. Also in step 58, the amplitudes of thedigitized and filtered down-converted in-phase signal 86 anddown-converted quadrature-phase signal 88 are determined at which acharacteristic of one sideband tone increases and that characteristic ofthe other three sideband tones decreases, as represented by thefrequency coefficients C₁₁, C₁₂, C₁₁ and C₁₂ of the four sideband tones.For an implementation in which reference signal 62 has the singlesideband tone at the particular frequency bin (0, 0, 1, 0 . . . 0, 0, 0)that results in evaluation signal 105 having the sideband tones at thevarious frequency bins (0, 0 . . . C₁₁ . . . C₁₂ . . . C₁₁ . . . C₁₂ . .. 0, 0, 0), the frequency coefficient that increases is C₁₁. In otherimplementations in which different frequency bins are used, thefrequency coefficient that increases is C₁₂.

The multiplication factors for multipliers 67-70 and 97-100 arecalculated by assuming a unity matrix Q that results in idealup-conversion and down-conversion. A complex data signal that istransmitted and received by transceiver 10 is expressed as:

x(t)=[

¦x(t)·a(t)¦·b(t)]_(BB)  (123).

Equation 123 is rewritten by separating the real and imaginarycomponents and including the unity matrix Q. FIG. 19 shows the resultingrewritten equation 124 representing the baseband components of a complexdata signal after up-conversion by a(t) and down-conversion by b(t),where ω_(a)=ω_(b). Because multiplication by the unity matrix Q resultsin ideal up-conversion and down-conversion, multiplication by theinverse of the actual matrix Q (Q¹⁻¹) will result in the unity matrixand compensate for any I/Q gain and phase mismatches. Thus, correctionof the I/Q impairments is achieved by multiplying the data signal by themultiplication factors corresponding to the gain coefficients of theinverse matrix Q⁻¹. FIG. 20 shows the coefficients of the inverse matrixQ⁻¹ that correspond to the multiplication factors of the multipliers67-70 and 97-100.

To determine the multiplication factors for multipliers 67-70 that areused to correct for the I/Q gain and phase mismatches of the quadraturemixer of transmitter 12, the quadrature mixer of receiver 13 is assumedto be ideal with impairments ε_(b)=φ_(b)=0. Then Q is solved for inequation 124 and the values of

{a(t)} and

{b(t)} are inserted from equations 79 and 80 and the values of

{b(t)} and

{b(t)} are inserted from equations 91 and 92. To determine themultiplication factors for multipliers 97-100 that are used to correctfor the I/Q gain and phase mismatches of the quadrature mixer ofreceiver 13, the impairments of transmitter 12 are set to zero, i.e.,ε_(a)=φ_(a)=0. Thus, when ε_(b)=φ_(b)=0 and the gain impairment ε_(a)and the phase mismatch φ_(a) are used in the calculation, the firstmultiplication factor of multiplier 67 is the coefficient at matrixposition 11 of FIG. 20; the second multiplication factor of multiplier68 is the coefficient at matrix position 12; the third multiplicationfactor of multiplier 69 is the coefficient at matrix position 21; andthe fourth multiplication factor of multiplier 70 is the coefficient atmatrix position 22. When ε_(a)=φ_(a)=0 and the gain impairment ε_(b) andthe phase mismatch φ_(b) are used in the calculation, the fifthmultiplication factor of multiplier 97 is the coefficient at matrixposition 11; the sixth multiplication factor of multiplier 98 is thecoefficient at matrix position 12; the seventh multiplication factor ofmultiplier 99 is the coefficient at matrix position 21; and the eighthmultiplication factor of multiplier 100 is the coefficient at matrixposition 22.

When ε_(b)=φ_(b)=0 and the gain impairment ε_(a) and the phase mismatchφ_(a) are used in the calculation, solving for Q in equation 124 resultsin equation 125 of FIG. 21. The inverse of Q is then calculated toarrive at the gain coefficients used by first correction circuit 22 tocorrect for the gain and phase impairments of transmitter 12. FIG. 22shows an equation 126 that describes the multiplication factors of thefour multipliers 67-70. For example, to determine the firstmultiplication factor for multiplier 67, the common multiplier ofequation 126 is multiplied by the matrix position 11, which equals(1−ε_(a)/2)·cos(φ_(a)/2).

To determine the multiplication factors for multipliers 97-100 that areused to correct for the I/Q gain and phase mismatches of the quadraturemixer of receiver 13, the quadrature mixer of transmitter 12 is assumedto be ideal with impairments ε_(a)=φ_(a)=0. Then, as for the transmittercorrection factors, Q is solved for in equation 124 and the values of

{a(t)},

{a(t)},

{b(t)} and

{b(t)} are inserted from equations 79, 80, 91 and 92.

When ε_(a)=φ_(a)=0 and the gain impairment ε_(b) and the phase mismatchφ_(b) are used in the calculation, solving for Q in equation 124 resultsin equation 127 of FIG. 23. The inverse of Q is then calculated toarrive at the gain coefficients used by second correction circuit 44 tocorrect for the gain and phase impairments of receiver 13. FIG. 24 showsan equation 128 that describes the multiplication factors of the fourmultipliers 97-100. For example, to determine the sixth multiplicationfactor for multiplier 98, the common multiplier of equation 128 ismultiplied by the matrix position 12, which equals−(1+ε_(b)/2)·sin(φ_(b)/2).

In summary, in the first phase of method 48 when transceiver 10 operatesin the estimation mode, reference signal 62 is generated and evaluationsignal 105 is observed. The complex frequency coefficients C₁₁, C₁₂, C₁₁and C₁₂ are determined by measuring evaluation signal 105. In a seriesof calculations performed in DSP 14, gain components K₁, K₁, K₁, andK¹⁻⁻ and phase components φ₁ and φ¹⁻ are calculated. Using the gaincomponents and the phase components, all four I/Q impairments (gainimpairments ε_(a) and ε_(b) and phase mismatches φ_(a) and φ_(b)) aredetermined at one time based on a single reference signal. The fourimpairments are used to calculate eight multiplication factors used inthe second phase of method 48 when transceiver is operating in thecorrection mode. The first four multiplication factors are used by themultipliers of first correction circuit 22 to correction the gainimpairment ε_(a) and phase impairment φ_(a) of transmitter 12, and thesecond four multiplication factors are used by the multipliers of secondcorrection circuit 44 to correction the gain impairment ε_(b) and phaseimpairment φ_(b) of receiver 13.

When data signals are transmitted in the correction mode, the I Qimpairments are corrected by pre-processing the data signals with firstcorrection circuit 22 to correct the gain and phase impairments oftransmitter 12 and by post-processing the data signals received withsecond correction circuit 44 to correct for the gain and phaseimpairments of receiver 13. For example, pre-processing is performedafter inverse Fourier transform calculation circuit 21 receives a datasignal 129 containing information transmitted by a user of the OFDMcommunication system. IFFT 21 generates I-phase Tx component 64 andQ-phase Tx component 65. I-phase Tx component 64 is received by firstmultiplier 67 and third multiplier 69, and Q-phase Tx component 65 isreceived by second multiplier 68 and fourth multiplier 70. In the secondphase of method 48 in which the four I/Q impairments are corrected,first correction circuit 22 does not merely passes through I-phasecomponent 64 and Q-phase component 65 unchanged. Instead, themultiplication factors calculated in the estimation mode determine howmuch of each of components 64 and 65 will be added to the othercomponent.

In a step 59, an attenuated component of I-phase Tx component 64 isgenerated using the first multiplication factor. DSP 14 sends a digitalcontrol signal 130 that includes the first multiplication factor tomultiplier 67. Multiplier 67 attenuates the amplitude of I-phase Txcomponent 64 by an amount corresponding to the first multiplicationfactor. In one embodiment, all of the multiplication factors cause themultipliers to attenuate the components 64 and 65. In anotherembodiment, the multiplication factors cause the multipliers either toamplify or to attenuate the components 64 and 65.

In a step 60, an attenuated component of Q-phase Tx component 65 isgenerated when multiplier 68 attenuates the amplitude of Q-phase Txcomponent 65 by an amount corresponding to the second multiplicationfactor.

In a step 61, the attenuated component of I-phase Tx component 64 isadded to the attenuated component of Q-phase Tx component 65. Firstadder 71 receives the outputs of first multiplier 67 and secondmultiplier 68 and generates a corrected I-phase Tx component 131.

In addition, multiplier 69 attenuates I-phase Tx component 64 by anamount corresponding to the third multiplication factor, and multiplier70 attenuates Q-phase Tx component 65 by an amount corresponding to thefourth multiplication factor. Second adder 72 receives the outputs ofthird multiplier 69 and fourth multiplier 70 and generates a correctedQ-phase Tx component 132. By adding a small amount of the quadraturephase component to the in-phase component and vice versa, both the phaseand the amplitude of each component is modified and thereby corrected.The data signal 129 is first pre-processed to compensate for the I/Qgain and phase impairments that are later introduced by the transmitquadrature mixer and is then up-converted and transmitted as correctedtransmission signal 133.

The correction mode also involves post-processing down-convertedreceived signals to compensate for the I/Q gain and phase impairmentsthat were introduced by the receive quadrature mixer. Second correctioncircuit 44 corrects received RF composite signals to compensate for theI/Q mismatches of the quadrature mixer of receiver 13. An RF receivesignal 134 is received onto antenna 18, passes through duplexer 17, isamplified by low noise amplifier 34, and is then down-converted by thequadrature mixer of receiver 13. The quadrature mixer outputsdown-converted in-phase signal 86 and down-converted quadrature-phasesignal 88. Signals 86 and 88 are then filtered and digitized. Digitizedand filtered in-phase signal 86 is received by fifth multiplier 97 andseventh multiplier 99, and digitized and filtered quadrature-phasesignal 88 is received by sixth multiplier 98 and eighth multiplier 100.The multiplication factors calculated in the estimation mode are thenused to govern how much of each of signal 86 and 88 will be added to theother signal by second correction circuit 44.

For example, an attenuated component of digitized and filtered in-phasesignal 86 is generated using the fifth multiplication factor. DSP 14sends a digital control signal 135 that includes the fifthmultiplication factor to multiplier 97. Multiplier 97 attenuates thedigitized and filtered in-phase signal 86 by an amount corresponding tothe fifth multiplication factor. In addition, an attenuated component ofdigitized and filtered down-converted quadrature-phase signal 88 isgenerated when multiplier 98 attenuates the amplitude of digitized andfiltered down-convened quadrature-phase signal 88 by an amountcorresponding to the sixth multiplication factor. Then, the attenuatedcomponent of signal 88 is added to the attenuated component of signal86. Third adder 101 receives the output of fifth multiplier 97 and sixthmultiplier 98 and generates a corrected I-phase Rx component 136.

In addition, multiplier 99 attenuates digitized and filtered in-phasesignal 86 by an amount corresponding to the seventh multiplicationfactor, and multiplier 100 attenuates digitized and filtereddown-convened quadrature-phase signal 88 by an amount corresponding tothe eighth multiplication factor. Fourth adder 102 receives the outputof seventh multiplier 99 and eighth multiplier 100 and generates acorrected Q-phase Rx component 137. Fourier transform calculationcircuit 45 then transforms corrected I-phase Rx component 136 andcorrected Q-phase Rx component 137 into a corrected baseband signal 138.The digital streams of corrected baseband signal 138 are then convertedinto symbols for subsequent digital signal processing. Second correctioncircuit 44 has removed the gain and phase impairments ε_(b) and φ_(b)from corrected baseband signal 138 that were introduced by thequadrature mixer of receiver 13. Thus, in the correction mode,transceiver 10 both pre-processes data signals to be transmitted andpost-processes received data signals to correct for the four I/Q gainand phase mismatches introduced by the quadrature mixers of thetransmitter and receiver. First correction circuit 22 pre-processes datasignal 129 before data signal 129 is up-converted and transmitted so asto correct for the gain impairment ε_(a) and the phase mismatch φ_(a)that will be introduced by transmitter 12. Second correction circuit 44post-processes RF receive signal 134 after signal 134 is received anddown-converted so as to correct for the gain impairment ε_(b) and thephase mismatch φ_(b) that were introduced by receiver 13. As indicatedby equation 107, when first correction circuit 22 and second correctioncircuit 44 compensate for the I/Q impairments by adding in-phase andquadrature-phase components to each other, the frequency coefficientC¹⁻¹ increases and the frequency coefficients C¹⁻¹, C¹⁻² and C¹⁻²decrease. In the ideal case where first correction circuit 22 and secondcorrection circuit 44 completely correct for the four I/Q impairments,the frequency coefficients C⁻¹, C¹⁻² and C¹⁻² have no amplitudes. Inthis ideal case, the diagram of FIG. 10 would exhibit only one peak, thepeak near frequency bin −40 corresponding to frequency coefficientsC¹⁻¹. Where the characteristics of the channel connection are not idealand where RF loop-back path 11 causes gain scaling and a time delayacross the channel connection, the impairments of the channel connectionimpact the frequency coefficients. Thus, impairments of the channelconnection change the phase of all sidebands equally and do not changethe relative magnitudes of the frequency coefficients.

In one embodiment, during normal operation while a user of the OFDMsystem is sending and receiving data signals, the operation oftransceiver 10 alternates between transmitting user data andre-calibrating the transmit and receive chains. After data signals havebeen transmitted and received for a predetermined amount of time,transceiver 10 enters its estimation mode. In one implementation,frequency synthesizer 36 changes the frequency ω_(b) of LO signal 47that is used in the estimation mode from the frequency used todown-convert received data signals. For example, the frequency ω_(b) ischanged to be within about 100 kilohertz of the frequency ω_(a)generated by transmit frequency synthesizer 28 and used to up-convertsignals to be transmitted. Transceiver 10 then generates referencesignal 62, evaluates evaluation signal 105, and determines themultiplication factors. Then after the transmit and receive chains havebeen re-calibrated, transceiver 10 alternates back to the correctionmode and transmits and receives data signals containing voice and datacommunications. In the correction mode, transceiver 10 again uses theoriginal frequency ω_(b) of LO signal 47 that is specified by thegovernmental spectrum licensing authority. In another embodiment,transceiver 10 enters its estimation mode only on powering up the mobilecommunication device that includes transceiver 10. After calibration atpowering up, transceiver operates in the correction mode.

FIG. 25 is a schematic block diagram of an embodiment of transceiver 10in which the correction circuits are located in analog RF transceiver IC19 instead of in digital baseband IC 20. Thus, the method ofsimultaneously estimating the gain and phase mismatches of both thetransmitter and the receiver corrects for the I/Q impairments in theanalog domain. First correction circuit 22 is located on analog RFtransceiver IC 19 and receives both an analog I-phase Tx componentoutput by first DAC 23 as well as an analog Q-phase Tx component outputby DAC 24. Attenuated components of I-phase Tx component 64 and Q-phaseTx component 65 are added to each other in the analog domain and outputas corrected I-phase Tx component 131 and corrected Q-phase Tx component132. Similarly, second correction circuit 44 is also located on analogRF transceiver IC 19 and receives filtered down-converted in-phasesignal 86 and filtered down-converted quadrature-phase signal 88. Secondcorrection circuit 44 then adds attenuated components of signals 86 and88 to each other and outputs corrected I-phase Rx component 136 andcorrected Q-phase Rx component 137. Corrected I-phase Rx component 136is then digitized by ADC 42, and corrected Q-phase Rx component 137 isdigitized by ADC 43.

The analog correction circuits of the embodiment of FIG. 25 arecontrolled by I/Q correction registers. Digital baseband IC 20communicates the multiplication factors and other correction informationacross SPI serial bus 15, through the I/Q correction registers, and tothe correction circuits. For example, FIG. 25 shows an I/Q correctionregister 139 that controls multipliers 97-100 of second correctioncircuit 44. The I/Q correction registers that control first correctioncircuit 22 are not shown in FIG. 25. In one aspect, multiplier 97 iscontrolled by correction signals received from I/Q correction register139. A control signal 140 is asserted when a digital one is written intothe eighth bit of I/Q correction register 139, and a control signal 141is asserted when a digital one is written into the seventh bit of I/Qcorrection register 139. In other aspects, more than two control signalsare used to set the multiplication factor of each multiplier. Digitalbaseband IC 20 sets the multiplication factor of multiplier 97 bycommunicating across SPI serial bus 15 and writing the digital values00, 01, 10 or 11 into the seventh and eighth bits of I/Q correctionregister 139.

FIG. 26 is a schematic block diagram of another embodiment oftransceiver 10 in which the I/Q impairments are not pre-processed andpost-processed, but rather are corrected at their source. The embodimentof FIG. 26 includes correction multipliers and correction shifterswithin the quadrature mixers that compensate for the gain mismatch andphase mismatches. Without correction, in-phase transmitter LO signal 73has a phase that is shifted ω_(a) from quadrature-phase transmitter LOsignal 75. In FIG. 26, the “90°−φ_(a)/2” in transmit phase shifter 29and the “+φ_(a)/2” above phase shifter 29 represent the phase mismatchof φ_(a) introduced by the quadrature mixer of transmitter 12. In oneembodiment, analog transmit phase shifter 29 is controlled by correctionsignals from an I/Q correction register. The phase shift generated bytransmit phase shifter 29 is changed such that the phase mismatch φ_(a)is eliminated. DSP 14 uses the phase mismatch φ_(a) obtained fromequation 119 to control transmit phase shifter 29 by writing bits of theI/Q correction register. In another embodiment, a correction shifter 142is added to the quadrature mixer in order to shift the phase ofquadrature-phase transmitter LO signal 75 by an amount −φ_(a) thatcorrects for the +φ_(a) phase mismatch introduced by the quadraturemixer.

In FIG. 26, multipliers 77 and 78 represent the gain mismatch of ε_(a)between up-converted in-phase signal 74 and up-convertedquadrature-phase signal 76. An analog, voltage-controlled correctionamplifier 143 is added to the quadrature mixer in order to attenuatesignal 74 by an amount −ε_(a)/2 that corrects for the +ε_(a)/2 gainmismatch introduced into signal 74 by the quadrature mixer. Similarly,an analog, voltage-controlled correction amplifier 144 is added to thequadrature mixer in order to amplify signal 76 by an amount +ε_(a)/2that corrects for the −ε_(a)/2 gain mismatch introduced into signal 76by the quadrature mixer. DSP 14 uses the gain mismatch ε_(a) obtainedfrom equation 115 in FIG. 17 to control the voltage-controlledamplifiers 143 and 144 by writing to bits of an I/Q correction register.In some implementations, the gain mismatch ε_(a) is a negative amount,such that amplifier 143 amplifies signal 74 and amplifier 144 attenuatessignal 76.

The embodiment of FIG. 26 also includes correction multipliers andcorrection shifters within the quadrature mixer of receiver 13. In oneembodiment, both quadrature mixers of the transmit and receive pathshave a single correction shifter that shifts the LO signal to one mixerby φ_(a) or φ_(b), such as correction shifter 142. In anotherembodiment, each quadrature mixer has two correction shifters, each ofwhich shifts the LO signal to one of the two mixers by φ/2. FIG. 26illustrates the double-shifter embodiment with regard to the quadraturemixer of receiver 13. A correction shifter 145 is added to the receivequadrature mixer in order to shift the phase of in-phase receiver LOsignal 85 by an amount −φ_(b)/2 that corrects for the +φ_(b)/2 phasemismatch introduced into signal 85 by the quadrature mixer. Similarly, acorrection shifter 146 is added to the receive quadrature mixer in orderto shift the phase of quadrature-phase receiver LO signal 87 by anamount +φ_(b)/2 that corrects for the −φ_(b)/2 phase mismatch introducedinto signal 87 by the quadrature mixer. In addition, a correctionamplifier 147 is added to the quadrature mixer in order to attenuatedown-converted in-phase signal 86 by an amount −ε_(a)/2 that correctsfor the +ε_(a)/2 gain mismatch introduced into signal 86 by thequadrature mixer. Similarly, a correction amplifier 148 is added to thequadrature mixer in order to amplify down-converted quadrature-phasesignal 88 by an amount +ε_(a)/2 that corrects for the −ε_(a)/2 gainmismatch introduced into signal 88 by the quadrature mixer.

The correction shifters and correction multipliers of the embodiment ofFIG. 26 are controlled by I Q correction registers. Digital baseband IC20 communicates correction information across SPI serial bus 15, throughthe I Q correction registers, and to the correction shifters andcorrection multipliers. The correction information includes the gainmismatches ε_(a) and ε_(b) obtained from equations 115-118 in FIG. 17and the phase mismatches φ_(a) and φ_(b) obtained from equations119-120. For example, FIG. 26 shows an I/Q correction register 149 thatcontrols correction shifters 145-146 and correction multipliers 147-148of the quadrature mixer of receiver 13. The I/Q correction registersthat control the correction shifters and correction multipliers of thequadrature mixer of transmitter 12 are not shown in FIG. 26. Thecorrection shifters and correction multipliers are controlled bycorrection signals received from I/Q correction register 149. Forexample, a control signal 150 is asserted when a digital one is writteninto the eighth bit of I/Q correction register 149, and a control signal151 is asserted when a digital one is written into the seventh bit ofI/Q correction register 149. In other examples, more than two controlsignals are used to set the attenuation amount for correction multiplier147. Digital baseband IC 20 sets the attenuation and amplificationamounts for correction multipliers 147-148 and the phase delays forcorrection shifters 145-146 by communicating across SPI serial bus 15and writing digital values into the eight bits of I/Q correctionregister 149.

In one or more exemplary embodiments, the functions described may beimplemented in hardware, software, firmware, or any combination thereof.If implemented in software, the functions may be stored on ortransmitted over as one or more instructions or code on acomputer-readable medium. Computer-readable media includes both computerstorage media and communication media including any medium thatfacilitates transfer of a computer program from one place to another. Astorage media may be any available media that can be accessed by acomputer. By way of example, and not limitation, such computer-readablemedia can comprise RAM, ROM, EEPROM. CD-ROM or other optical diskstorage, magnetic disk storage or other magnetic storage devices, or anyother medium that can be used to carry or store desired program code inthe form of instructions or data structures and that can be accessed bya computer. Also, any connection is properly termed a computer-readablemedium. For example, if the software is transmitted from a website,server, or other remote source using a coaxial cable, fiber optic cable,twisted pair, digital subscriber line (DSL), or wireless technologiessuch as infrared, radio, and microwave, then the coaxial cable, fiberoptic cable, twisted pair. DSL, or wireless technologies such asinfrared, radio, and microwave are included in the definition of medium.Disk and disc, as used herein, includes compact disc (CD), laser disc,optical disc, digital versatile disc (DVD), floppy disk and blu-ray discwhere disks usually reproduce data magnetically, while discs reproducedata optically with lasers. Combinations of the above should also beincluded within the scope of computer-readable media.

Although a transceiver that determines the gain and phase mismatches ofboth the transmit and receive paths after observing a single evaluationsignal has been described in connection with certain specificembodiments for instructional purposes, the transceiver is not limitedthereto. For example, transceiver 10 is described as performing thecalculation of the estimation mode in a digital signal processor. Inother embodiments, the calculations are performed in other parts of thedigital baseband IC. For example, the calculations are performed in anembedded microcontroller or in embedded programmable logic. Transceiver10 is described as controlling the correction circuits, correctionshifters and correction multipliers using control signals sent fromregisters. In other embodiments, control signals are sent directly fromthe digital baseband IC. The method of simultaneously calculating allfour I/Q impairments is described as being performed in a mobilecommunication device. In other embodiments, the method is performed in abase station that receives signals from and transmits signals to mobilecommunication devices. The previous description of the disclosedembodiments is provided to enable any person skilled in the art to makeor use a stepped gain mixer. Various modifications to these embodimentswill be readily apparent to those skilled in the art, and the genericprinciples defined herein may be applied to other embodiments withoutdeparting from the spirit or scope of disclosed subject matter.Accordingly, the disclosed method for determining the gain and phasemismatches of both the transmit and receive paths after observing asingle evaluation signal is not intended to be limited to theembodiments shown herein but is to be accorded the widest scopeconsistent with the principles and novel features disclosed herein.

1. A method comprising: receiving a reference signal into a quadraturemixer of a transmit path, wherein the transmit path exhibits a gainmismatch and a phase mismatch; generating an up-converted RF signalbased on the reference signal; transmitting the up-converted RF signalvia an RF loop-back path to a quadrature mixer of a receive path,wherein the receive path exhibits a gain mismatch and a phase mismatch:generating a down-converted evaluation signal based on the up-convertedRF signal; and determining the gain mismatch and the phase mismatch ofthe transmit path and the gain mismatch and the phase mismatch of thereceive path based on the down-converted evaluation signal.
 2. Themethod of claim 1, wherein the quadrature mixer of the transmit path ispart of a communication device that communicates based on orthogonalfrequency-division multiplexing (OFDM) and frequency division duplexing(FDD).
 3. The method of claim 2, wherein the communication deviceperforms regular transmission of data signals, and wherein thedetermining the gain mismatch and the phase mismatch of the transmitpath and the gain mismatch and the phase mismatch of the receive path isperformed without using hardware other than the RF loop-back path thatis not otherwise used in the regular transmission of data signals. 4.The method of claim 1, wherein the evaluation signal has a firstsideband tone with a first amplitude, a second sideband tone with asecond amplitude, a third sideband tone with a third amplitude and afourth sideband tone with a fourth amplitude, and wherein the gainmismatch and the phase mismatch of the transmit path and the gainmismatch and the phase of the receive path are determined based on thefirst amplitude, the second amplitude, the third amplitude and thefourth amplitude.
 5. A method comprising: generating a reference signal;transforming the reference signal into an in-phase Tx component and aquadrature-phase Tx component; generating an up-converted in-phasesignal by mixing the in-phase Tx component with an in-phase transmitteroscillator signal; generating an up-converted quadrature-phase signal bymixing the quadrature-phase Tx component with a quadrature-phasetransmitter oscillator signal; generating a composite RF signal byadding the up-converted in-phase signal and the up-convertedquadrature-phase signal; generating a down-converted in-phase signal bymixing the composite RF signal with an in-phase receiver oscillatorsignal; generating a down-converted quadrature-phase signal by mixingthe composite RF signal with a quadrature-phase receiver oscillatorsignal; transforming the down-converted in-phase signal and thedown-converted quadrature-phase signal into an evaluation signal,wherein the evaluation signal has a first sideband tone with a firstcharacteristic, a second sideband tone with a second characteristic, athird sideband tone with a third characteristic and a fourth sidebandtone with a fourth characteristic; determining the first characteristic,the second characteristic, the third characteristic and the fourthcharacteristic; and determining an amplitude of the down-convertedin-phase signal and an amplitude of the down-converted quadrature-phasesignal at which the first characteristic increases and each of thesecond characteristic, the third characteristic and the fourthcharacteristic decreases.
 6. The method of claim 5, wherein the in-phasetransmitter oscillator signal has a first frequency and the in-phasereceiver oscillator signal has a second frequency, and wherein the firstfrequency differs from the second frequency.
 7. The method of claim 5,wherein the characteristic is amplitude.
 8. The method of claim 5,wherein the first characteristic is a combination of amplitude and phaseof the first sideband tone.
 9. The method of claim 8, wherein the firstcharacteristic is defined by a complex Fourier coefficient.
 10. Themethod of claim 5, wherein the reference signal is generated infrequency domain and has an amplitude in only one frequency bin.
 11. Themethod of claim 5, wherein the reference signal has a single sidebandtone.
 12. The method of claim 5, further comprising: generating anattenuated component of the down-converted in-phase signal; generatingan attenuated component of the down-converted quadrature-phase signal;and adding the attenuated component of the down-convertedquadrature-phase signal to the attenuated component of thedown-converted in-phase signal.
 13. The method of claim 5, wherein thegenerating the composite RF signal is performed on an analog integratedcircuit, and wherein the transforming the down-converted in-phase signaland the down-converted quadrature-phase signal is performed on a digitalintegrated circuit.
 14. The method of claim 5, wherein the generatingthe composite RF signal is performed in a time domain, and u herein thetransforming the down-corn cried in-phase signal and the down-convertedquadrature-phase signal is performed in a frequency domain.
 15. Themethod of claim 5, wherein the in-phase Tx component has an amplitudeand wherein the quadrature-phase Tx component has an amplitude, furthercomprising: adjusting the amplitude of the in-phase Tx component and theamplitude of the quadrature-phase Tx component; and adding the adjustedin-phase Tx component to the adjusted quadrature-phase Tx component. 16.A circuit comprising: a first multiplier that receives a down-convertedin-phase signal and generates a first attenuated component signal havingan amplitude; a second multiplier that receives a down-convertedquadrature-phase signal and generates a second attenuated componentsignal having an amplitude, wherein a data signal with a single sidebandtone is transmitted and used to generate the down-converted in-phasesignal and the down-converted quadrature-phase signal; a thirdmultiplier that receives the down-converted in-phase signal andgenerates a third attenuated component signal having an amplitude; afourth multiplier that receives the down-converted quadrature-phasesignal and generates a fourth attenuated component signal having anamplitude; a first adder that generates an in-phase component signal byadding the second attenuated component signal to the first attenuatedcomponent signal; a second adder that generates a quadrature-phasecomponent signal by adding the third attenuated component signal to thefourth attenuated component signal; and a Fourier transform calculationcircuit that generates a corrected signal using the in-phase componentsignal and the quadrature-phase component signal, wherein the correctedsignal has a first sideband tone with a first characteristic, a secondsideband tone with a second characteristic, a third sideband tone with athird characteristic and a fourth sideband tone with a fourthcharacteristic, and wherein the first multiplier adjusts the amplitudeof the first attenuated component signal, the second multiplier adjuststhe amplitude of the second attenuated component signal, the thirdmultiplier adjusts the amplitude of the third attenuated componentsignal and the fourth multiplier adjusts the amplitude of the fourthattenuated component signal all such that the first characteristicincreases and the second characteristic, the third characteristic andthe fourth characteristic decrease.
 17. The circuit of claim 16, whereinthe corrected signal is an evaluation signal.
 18. The circuit of claim16, wherein the characteristic is amplitude.
 19. The circuit of claim16, wherein the first characteristic is defined by a complex Fouriercoefficient.
 20. The circuit of claim 16, wherein the down-convertedin-phase signal is filtered and digitized before being received by thefirst multiplier and by the third multiplier, and wherein thedown-converted quadrature-phase signal is filtered and digitized beforebeing received by the second multiplier and by the fourth multiplier.21. The circuit of claim 16, wherein the down-converted in-phase signalreceived by the first multiplier and the third multiplier is a digitizeddown-converted in-phase signal, and wherein the down-convertedquadrature-phase signal received by the second multiplier and the fourthmultiplier is a digitized down-converted quadrature-phase signal,further comprising: a first analog-to-digital converter that receives ananalog down-converted in-phase signal and outputs the digitizeddown-converted in-phase signal; and a second analog-to-digital converterthat receives an analog down-converted quadrature-phase signal andoutputs the digitized down-converted quadrature-phase signal.
 22. Thecircuit of claim 21, wherein the analog down-converted in-phase signalis filtered before being received by the first analog-to-digitalconverter.
 23. The circuit of claim 16, further comprising: an inverseFourier transform calculation circuit that receives the data signal andgenerates an in-phase component and a quadrature-phase component,wherein the in-phase component and the quadrature-phase component areused to generate the down-converted in-phase signal and thedown-converted quadrature signal.
 24. The circuit of claim 16, whereinthe circuit is part of a communication device that communicates based onorthogonal frequency-division multiplexing (OFDM) and frequency divisionduplexing (FDD).
 25. A circuit that operates in a first mode and in asecond mode, comprising: an inverse Fourier transform calculationcircuit that outputs an I-phase Tx component and a Q-phase Tx component;a first correction circuit having a first multiplier and a thirdmultiplier that receive the I-phase Tx component and having a secondmultiplier and a fourth multiplier that receive the Q-phase Txcomponent; a second correction circuit having a fifth multiplier and aseventh multiplier that receive a down-converted in-phase signal andhaving a sixth multiplier and an eighth multiplier that receive adown-converted quadrature-phase signal, wherein the second correctioncircuit generates an I-phase Rx component and a Q-phase Rx component: aFourier transform calculation circuit that receives the I-phase Rxcomponent and the Q-phase Rx component; and a digital signal processor,wherein in the first mode the inverse Fourier transform calculationcircuit receives a reference signal and the Fourier transformcalculation circuit generates an evaluation signal, wherein theevaluation signal has a first sideband tone with a first characteristic,a second sideband tone with a second characteristic, a third sidebandlone with a third characteristic and a fourth sideband tone with afourth characteristic, wherein the digital signal processor uses theevaluation signal to determine the first characteristic, the secondcharacteristic, the third characteristic and the fourth characteristic,wherein the digital signal processor determines a first multiplicationfactor and a third multiplication factor by which the first and thethird multipliers attenuate the I-phase Tx component, a secondmultiplication factor and a fourth multiplication factor by which thesecond and the fourth multipliers attenuate the Q-phase Tx component, afifth multiplication factor and a seventh multiplication factor by whichthe fifth and the seventh multipliers attenuate the down-convenedin-phase signal, and a sixth multiplication factor and an eighthmultiplication factor by which the sixth and the eighth multipliersattenuate the down-converted quadrature-phase signal, and wherein thedigital signal processor determines the first, second, third, fourth,fifth, sixth, seventh and eighth multiplication factors such that thefirst characteristic increases and each of the second characteristic,the third characteristic and the fourth characteristic decreases. 26.The circuit of claim 25, wherein in the second mode the inverse Fouriertransform calculation circuit receives a data signal and the Fouriertransform calculation circuit outputs a corrected baseband signal, andwherein the circuit that operates in a first mode and in a second modeuses the first, second, third, fourth, fifth, sixth, seventh and eighthmultiplication factors determined in the first mode to generate theI-phase Rx component and the Q-phase Rx component received by theFourier transform calculation circuit in the second mode.
 27. Thecircuit of claim 25, wherein the characteristic is amplitude.
 28. Thecircuit of claim 25, wherein the first characteristic is a combinationof amplitude and phase of the first sideband tone.
 29. The circuit ofclaim 25, wherein the first characteristic is defined by a complexFourier coefficient.
 30. The circuit of claim 25, wherein thedown-converted in-phase signal is filtered and digitized before beingreceived by the fifth multiplier and by the seventh multiplier, andwherein the down-converted quadrature-phase signal is filtered anddigitized before being received by the sixth multiplier and by theeighth multiplier.
 31. The circuit of claim 25, wherein the firstcorrection circuit further comprises a first adder and a second adder,wherein the first adder is coupled to the first multiplier and to thethird multiplier, wherein the second adder is coupled to the secondmultiplier and to the fourth multiplier, wherein the second correctioncircuit further comprises a third adder and a fourth adder, wherein thethird adder is coupled to the fifth multiplier and to the sixthmultiplier, and wherein the fourth adder is coupled to the seventhmultiplier and to the eighth multiplier.
 32. The circuit of claim 25,wherein the first correction circuit outputs I and Q transmitcomponents, and wherein the second correction circuit receives I and Qreceive components, further comprising: a transmit quadrature mixer thatreceives the I and Q transmit components and outputs a composite RFsignal; a receive quadrature mixer that receives the composite RF signaland outputs the I and Q receive components; and an RF loop-back pathover which the composite RF signal is conveyed from the transmitquadrature mixer to the receive quadrature mixer.
 33. The circuit ofclaim 31, wherein the circuit is part of a digital baseband integratedcircuit.
 34. A processor-readable medium for storing instructionsoperable in a wireless device to: generate a reference signal; transformthe reference signal into an in-phase Tx component and aquadrature-phase Tx component, wherein an up-converted in-phase signalis generated by mixing the in-phase Tx component with an in-phasetransmitter oscillator signal, wherein an up-converted quadrature-phasesignal is generated by mixing the quadrature-phase Tx component with aquadrature-phase transmitter oscillator signal, wherein a composite RFsignal is generated by adding the up-converted in-phase signal and theup-converted quadrature-phase signal, wherein a down-converted in-phasesignal is generated by mixing the composite RF signal with an in-phasereceiver oscillator signal, and wherein a down-convertedquadrature-phase signal is generated by mixing the composite RF signalwith a quadrature-phase receiver oscillator signal; transform thedown-converted in-phase signal and the down-converted quadrature-phasesignal into an evaluation signal, wherein the evaluation signal has afirst sideband lone with a first characteristic, a second sideband tonewith a second characteristic, a third sideband tone with a thirdcharacteristic and a fourth sideband lone with a fourth characteristic;determine the first characteristic, the second characteristic, the thirdcharacteristic and the fourth characteristic; and determine an amplitudeof the down-converted in-phase signal and an amplitude of thedown-converted quadrature-phase signal at which the first characteristicincreases and each of the second characteristic, the thirdcharacteristic and the fourth characteristic decreases.
 35. Theprocessor-readable medium of claim 34, wherein the in-phase transmitteroscillator signal has a first frequency and the in-phase receiveroscillator signal has a second frequency, and wherein the firstfrequency differs from the second frequency.
 36. The processor-readablemedium of claim 34, wherein the first characteristic is a combination ofamplitude and phase of the first sideband tone.
 37. Theprocessor-readable medium of claim 34, wherein the reference signal isgenerated in frequency domain and has an amplitude in only one frequencybin.
 38. The processor-readable medium of claim 34, wherein thereference signal has a single sideband tone.
 39. The processor-readablemedium of claim 34, and further for storing instructions operable in thewireless device to: adjust an amplitude of the in-phase Tx component andan amplitude of the quadrature-phase Tx component; and add the adjustedin-phase Tx component to the adjusted quadrature-phase Tx component. 40.A circuit comprising: a first correction circuit that corrects for bothan up-conversion phase mismatch and an up-conversion gain mismatchbetween an up-convened in-phase signal and an up-convertedquadrature-phase signal, wherein the first correction circuit multipliesan I-phase Tx component by a first multiplication factor and by a thirdmultiplication factor, and wherein the first correction circuitmultiplies a Q-phase Tx component by a second multiplication factor andby a fourth multiplication factor; a second correction circuit thatcorrects for both a down-conversion phase mismatch and a down-conversiongain mismatch between a down-converted in-phase signal and adown-converted quadrature-phase signal, wherein the second correctioncircuit multiplies the down-converted in-phase signal by fifthmultiplication factor and by a seventh multiplication factor, whereinthe second correction circuit multiplies the down-convertedquadrature-phase signal by sixth multiplication factor and by an eighthmultiplication factor; and means for determining the first, second,third, fourth, fifth, sixth, seventh and eighth multiplication factorsbased on evaluating a single evaluation signal.
 41. The circuit of claim40, wherein the single evaluation signal is generated from a singlereference signal, and wherein the single reference signal has a singlesideband tone.
 42. The circuit of claim 40, wherein the first correctioncircuit comprises a first adder and a second adder, wherein the firstadder is coupled to a first multiplier and to a third multiplier,wherein the second adder is coupled to a second multiplier and to afourth multiplier, wherein the second correction circuit comprises athird adder and a fourth adder, wherein the third adder is coupled to afifth multiplier and to a sixth multiplier, wherein the fourth adder iscoupled to a seventh multiplier and to an eighth multiplier, and whereinthe first, second, third, fourth, fifth, sixth, seventh and eighthmultipliers apply the first, second, third, fourth, fifth, sixth,seventh and eighth multiplication factors, respectively.
 43. The circuitof claim 40, wherein the circuit is part of a communication device thatcommunicates based on orthogonal frequency-division multiplexing (OFDM)and frequency division duplexing (FDD).